Hybrid MU-MIMO spatial mapping using both explicit sounding and crosstalk tracking in a wireless local area network

ABSTRACT

A wireless access point supporting a wireless local area network (WLAN) including multiple-input multiple-output (MIMO) communications with associated stations, is disclosed with hardware processing circuitry to perform hybrid spatial mapping operations for multi-user (MU) MIMO downlinks including: explicit channel sounding and crosstalk damping circuits. The explicit channel sounding circuit determines from communication channel feedback, a MU-MIMO precode matrix “Q” for spatially mapping a transmission of an MU-MIMO downlink communication packet to corresponding ones of a group of stations. The crosstalk damping circuit is responsive to crosstalk feedback from at least one of the stations in the MU-MIMO group to incrementally adjust elements of the precode matrix “Q” in a direction which reduces the amount of crosstalk; and to spatially map subsequently transmitted MU-MIMO downlink communication packets with the adjusted precode matrix “Q′”. A complementary wireless station is disclosed.

CROSS REFERENCE TO RELATED APPLICATION

This application claims the benefit of prior filed ProvisionalApplication No. 62/190,648 filed on Jul. 9, 2015 entitled “MU Precodingfor Changing Channels” which is incorporated herein by reference in itsentirety as if fully set forth herein.

BACKGROUND OF THE INVENTION 1. Field of Invention

The field of the present invention relates in general to wireless localarea networks (WLAN) including wireless access points (WAP) and wirelessstations and methods for spatial mapping of multi-user communications onsame.

2. Description of the Related Art

Home and office networks, a.k.a. wireless local area networks (WLAN) areestablished and serviced using a device called a Wireless Access Point(WAP). The WAP may include a router. The WAP wirelessly couples all thedevices of the home network, e.g. wireless stations such as: computers,printers, televisions, digital video (DVD) players, security cameras andsmoke detectors to one another and to the Cable or Subscriber Linethrough which Internet, video, and television is delivered to the home.Most WAPs implement the IEEE 802.11 standard which is a contention basedstandard for handling communications among multiple competing devicesfor a shared wireless communication medium on a selected one of aplurality of communication channels. The frequency range of eachcommunication channel is specified in the corresponding one of the IEEE802.11 protocols being implemented, e.g. “a”, “b”, “g”, “n”, “ac”, “ad”,“ax”. Communications follow a hub and spoke model with a WAP at the huband the spokes corresponding to the wireless links to each ‘client’device.

After selection of a single communication channel for the associatedhome network, access to the shared communication channel relies on amultiple access methodology identified as Collision Sense MultipleAccess (CSMA). CSMA is a distributed random access methodology firstintroduced for home wired networks such as Ethernet for sharing a singlecommunication medium, by having a contending communication link back offand retry access to the line if a collision is detected, i.e. if thewireless medium is in use.

Communications on the single communication medium are identified as“simplex” meaning, one communication stream from a single source node toone or more target nodes at one time, with all remaining nodes capableof “listening” to the subject transmission. To confirm arrival of eachcommunication packet, the target node is required to send back anacknowledgment, a.k.a. “ACK” packet to the source. Absent the receipt ofthe ACK packet the source will retransmit the unacknowledged data untilan acknowledgement is received, or a time-out is reached.

Where communications are from one source to more than one targetconcurrently, and where further the data for each target is discretefrom that destined for other targets, the communication is identified asmulti-user (MU) multiple-input multiple-output (MIMO) communication.This communication capability requires complex signal processing andmultiple antennas on at least the transmitting device, e.g. the wirelessaccess point (WAP). MU-MIMO capability was first introduced into theIEEE 802.11 specification in the “ac” standard and is intended toincrease WAP throughput by allowing the MIMO capabilities of the WAP tobe exploited efficiently by downlinking the WAP to multiple stations atonce via an MU-MIMO downlink communication packet having a payloadportion containing discrete individual communication packets for each ofthe targeted stations in the MU-MIMO group. The complex signalprocessing required to support this capability provides spatialseparation of each downlink packet transmitted by the WAP so that thediscrete payload portions of each packet arrive at each targeted stationin the MU-MIMO group without interference from the portion of eachpacket destined for others of the stations in the MU-MIMO group. Aprecode matrix input to a spatial mapper achieves this outcome. Theprecode matrix itself requires frequent interruptions of the MU-MIMOdownlink communications, in order for each station in the group to beexplicitly sounded, and further to provide feedback to the WAP as to thecommunication channel determined by each station in response to theexplicit sounding. Typically an MU-MIMO downlink will required ten ormore explicit soundings per second, i.e. at least 1 explicit soundingevery 100 ms. MU-MIMO is particularly appropriate or effective when thestations chosen for a MU-MIMO group have capabilities and correspondingdata consumption requirements which do not individually present a burdento the WAP. A typical MU-MIMO target station group would have a fractionof the antennas of the WAP and relatively low individual dataconsumption requirements.

What is needed is are improvements in MIMO communication capabilities onresidential/business WLAN.

SUMMARY OF THE INVENTION

The present invention provides a method and apparatus for reducing theoverhead associated with frequent explicit channel soundings formulti-user (MU) multiple-input multiple-output (MIMO) downlinkcommunication packet transmissions on a wireless local area network(WLAN) by use of crosstalk feedback from one or more of the group ofstations targeted for the MU-MIMO downlink communication packets duringthe course of such downlink communications.

In an embodiment of the invention a wireless access point (WAP) having aplurality of antennas and supporting a wireless local area network(WLAN) including multiple-input multiple-output (MIMO) communicationswith associated stations on a selected one of a plurality ofcommunication channels, is disclosed with hardware processing circuitryto perform hybrid spatial mapping operations for multi-user (MU) MIMOdownlinks including in such circuitry an explicit channel soundingcircuit and a crosstalk damping circuit. The explicit channel soundingcircuit determines from communication channel feedback, a multi-user(MU) MIMO precode matrix “Q” for spatially mapping a transmission ofdiscrete portions of a payload of an MU-MIMO downlink communicationpacket to achieve crosstalk-free reception at the corresponding ones ofa group of the associated stations targeted by the MU-MIMO downlink. Thecrosstalk damping circuit is responsive to crosstalk feedback from atleast one of the stations in the MU-MIMO group to incrementally adjustelements of the precode matrix “Q” in a direction which reduces theamount of crosstalk at the receivers; and to spatially map subsequentlytransmitted MU-MIMO downlink communication packets with the adjustedprecode matrix “Q′”, thereby improving downlink communications betweenexplicit soundings.

In another embodiment of the invention a complementary wireless stationis disclosed with hardware processing circuitry to perform hybridspatial mapping feedback operations for multi-user (MU) multiple-inputmultiple-output (MIMO) downlinks from the WAP to a group of associatedstations, and including in such circuitry a channel estimation circuitand a crosstalk estimation circuit. The channel estimation circuit isresponsive to an explicit sounding received from the WAP to determine acommunication channel associated therewith and to transmit feedback tothe WAP as to the determined communication channel. The crosstalkestimation circuit is responsive to receipt of a MU-MIMO downlinkcommunication packet from the WAP to determine an amount of crosstalkfrom portions of the downlink MU-MIMO communication packet targeted forother stations in the group and to transmit feedback to the WAP as tothe determined crosstalk.

Associated methods are also claimed.

BRIEF DESCRIPTION OF THE DRAWINGS

These and other features and advantages of the present invention willbecome more apparent to those skilled in the art from the followingdetailed description in conjunction with the appended drawings in which:

FIG. 1 is a timeline of a Prior Art MU-MIMO downlink showing soundingoverhead and downlink portions;

FIGS. 2A-2D are a representative Prior Art channel graph, a Prior ArtWLAN timing diagram including soundings, a Prior Art detailed explicitsounding timing diagram, and a Prior Art packet diagram of a WLANpackets including the sounding field.

FIG. 3 is a timeline of a MU-MIMO downlink in accordance with anembodiment of the invention incorporating hybrid spatial mapping usingboth explicit sounding and crosstalk tracking;

FIG. 4A is a graphs of communication channel variations as a function oftime;

FIGS. 4B-D are graphs of communication channel variations as a functionof time with and without explicit soundings;

FIG. 4E is a graph of explicit sounding overhead as a function ofmodulation and code schema (MCS) used to transmit sounding feedback;

FIGS. 4F-G are graphs of communication channel variations as a functionof time with the hybrid spatial mapping of an embodiment of theinvention;

FIG. 5 is a hardware block diagram of a WAP and two stations configuredto support the hybrid spatial mapping for MU-MIMO downlinks of anembodiment of the current invention;

FIG. 6 is a detailed circuit diagram of hybrid spatial mapping circuitsin the WAP and Stations as shown in FIG. 5, in accordance with anembodiment of the current invention; and

FIG. 7 is a process flow diagram of processes associated with hybridspatial mapping for MU-MIMO downlinks on both the WAP and stations inaccordance with an embodiment of the invention.

DETAILED DESCRIPTION OF TH EMBODIMENTS

FIG. 1 is a timeline of a Prior Art MU-MIMO downlink showing soundingoverhead and downlink portions. The timeline shows a wireless accesspoint (WAP) 100 and associated stations 102A-B of a residential 150wireless local area network (WLAN) during various phases of operationassociated with an MU-MIMO downlink. Each packet in an MU-MIMO downlinkhas a payload portion which contains distinct blocks of data eachdestined for a corresponding one of the stations in the MU-MIMO downlinkgroup. An MU-MIMO packet allows transmission of discrete payload data totwo or more receiving stations at the same time, thus increasing WLANthroughput. Ideally, each station in the group receives a common MU-MIMOpacket header, and its own distinct block of data. The distinct blocksof data in the MU-MIMO packet payload are said to be spatially mappedduring transmission so as to arrive at each corresponding one of thestations, without interference, a.k.a. crosstalk, from the distinctblocks of data destined for other stations in the group. Spatial mappinginvolves complex signal processing using the multiple antennas of theWAP and a precoding matrix to spatially separate the distinct blocks ofdata in the MU-MIMO packet payload among the targeted MU-MIMO group ofstations. The determination of the precode matrix, requires detailedchannel information feedback from each of the targeted stations inresponse to an explicit sounding. The feedback from each station as toits communication channel with the WAP is aggregated by the WAP and usedto determine the precode matrix for transmitting subsequent MU-MIMOdownlink packets. Minute changes in the communication channels blur thespatial separation so as to require frequent explicit soundings, e.g.ten per second, to recalculate the precode matrix. No data istransmitted during these explicit soundings, thus they are characterizedas overhead required to support MU-MIMO downlinks.

The timeline in FIG. 1 includes an overhead portion including thesounding and feedback phases “A” and “B” respectively of an Explicitsounding over a representative interval, e.g. 10 milliseconds. Theexplicit sounding is initiated at time to with the soundingtransmissions 108 by the WAP 100. The sounding is conducted on aselected one of the communication channels available to the WLAN. Thesounding phase is initiated with a null data packet announcement (NDPA)(not shown) which announces the group of target stations selected forthe MU-MIMO sounding and subsequent downlink. This announcement packetis followed by a null data packet (NDP) 110. Both the NDPA and the NDPpackets are sent using an isotropic RF signal strength 108. Neither theNDPA nor the NDP contains any user data. These packets are explicitlydedicated to soliciting a sounding of the link channel. The header 110Aof the NDP contains a Very High Throughput Long Training Field, a.k.a.VHT-LTF, which the receiving stations targeted for the MU-MIMO downlink,i.e. stations 102A-B use to characterize the respective link channelsH_(a) and H_(b). Each receiving station 102A-B determines the respectivelink channel H_(n), where “n” is the sub-channel or tone index, for eachof the OFDM tones or sub-channels of the selected channel. Each of thetargeted stations convert this channel information into feedback in theform of: a beamsteering matrix “V” and a per tone Signal-to-Noise Matrix“SNR”. The targeted stations 102A-B will then over the interval t₁-t₂,send the respective feedback packets 114, 112 including in the payloadportion thereof, the corresponding “V” and “SNR” matrices. If thesounding feedback conforms to the IEEE 802.11ac standard, each feedbackpacket 112, 114 will include the V and SNR matrices 112B, 114Brespectively for each sub-channel or tone of the correspondingWAP-station sounding. The WAP 100 aggregates this received feedback anddetermines a precoding matrix “Q” 118 therefrom.

The next portion of the prior art MU-MIMO timeline shown in FIG. 1 isthe downlink portion including the downlink transmission and receiptacknowledgement phases “C” and “D” respectively of an MU-MIMO downlinkover a representative interval, e.g. 90 milliseconds. The precodingmatrix “Q” 118 for each tone or subchannel, a.k.a. the beamsteeringmatrices, are starting at time t₂, used for spatially mapping thetransmitted MU-MIMO packet to the stations 102A-B. The MU-MIMO packet124 includes a header 124A and a payload portion. The spatial mapping ofthe payload portion of the MU-MIMO packet 124 allows the distinct blockof data for station 102A and the distinct block of data for station 102Bto arrive at their respective target stations without crosstalk orinterference from the other block of data. The precoding applied by thespatial mapper to the transmitted payload produces the distinctlyanisotropic RF signal strength multi-lobe pattern 120A-120B. At time t₃each station receiving the MU-MEMO packet 124 acknowledges receipt ofthe packet in an “ACK” packet. In this example where the MU-MIMO groupcomprises two stations 102A and 102B, each of which separatelyacknowledge receipt of the packet in ACK packets 132, 130 respectively.The WAP will resend any packet for which it has not received a receiptacknowledgement. Since slight changes in the respective communicationchannels are expected the soundings are frequently repeated, e.g. 100millisecond intervals in other words ten times per second, with theprecode matrix recalculated each time. Thus the overhead related tomaintaining the MU-MIMO downlink is 10% or 10 milliseconds out of every100 millisecond interval.

FIGS. 2A-2D are a representative Prior Art channel graph, a Prior ArtWLAN timing diagram including soundings, a Prior Art detailed explicitsounding timing diagram, and a Prior Art packet diagram of a WLANpackets including the sounding field.

FIG. 2A is a representative Prior Art channel graph in which the x and yaxis dimension frequency vs. power respectively for the eight 20 MHzcommunication channels, 200-214. Each communication channel may beselected individually to support a wireless local area network (WLAN).Alternately more than one of the 20 Mhz channels can be aggregated invarious combinations to form a 40 Mhz, 80 Mhz or 160 Mhz Aggregatechannel to support WLAN communications. Each 20 Mhz communicationchannel is orthogonal frequency division multiplexed (OFDM), i.e.divided into sub-channels or tones. Each 20 Mhz channel has 56independently modulated sub-carriers or tones. Communication channel 200has sub-channels a.k.a. tones, e.g. sub-channels 200A, 200B. Thischannel layout corresponds to that specified in IEEE 802.11ac forexample.

FIG. 2B is a representative Prior Art WLAN timing diagram includingsoundings. A typical operation of a WLAN includes beacon frames 220A-Esent periodically, e.g. at 100 ms intervals t₀-t₅. In a representativebeacon interval, t₀-t₁ the following sub-intervals are shown: acontention free interval 224A, a sounding interval 226A and a contentioninterval 228A. During the contention free interval the WAP sendsdownlink user data communications sequentially to one or more of thestations in the WLAN. During the sounding interval one or moredownstream links are probed to determine the channel characteristicsthereof and using the CSI in the feedback from the sounding thebeamforming matrix for each link subject to the sounding is determined.The soundings are conducted on a per link basis. The sounding feedbackis different for each link. During the contention based interval carriersense multiple access (CSMA) is used as a medium access control (MAC)methodology to allow any station to seize control of the channel andsend uplink user data communications thereon to the WAP. In the nextbeacon interval t₁-t₂ there is a contention free interval 224B and acontention interval 228B. In the next beacon interval t₂-t₃ there is acontention free interval 224C, including one or more soundings 226C, anda contention interval 228C.

FIG. 2C is a Prior Art detailed explicit sounding timing diagram for anMU-MIMO downlink showing a detailed view of a representative sounding226A as shown in FIG. 2B. An Explicit sounding of the link channelsbetween the WAP and station 108 and the WAP and station 112 are shown.The packets 240, 242, 244, 246, 248 are all associated with thesounding. All packets including those associated with the sounding,include a header portion shown in crosshatch and referenced with the “A”suffix. Following the sounding, downlink MU-MIMO packet transmissionsresume, and user data, e.g. MU-MIMO packet 250, is sent to the targetedgroup of stations using the precoding matrix derived from the channelsounding feedback.

The explicit sounding provided for in the IEEE 802.11ac standard allowsthe receiving station(s) to assist the transmitting WAP to steersubsequent user data communications toward the station(s) using thebeamforming matrix(s) provided by the station(s) in response to theexplicit channel sounding initiated by the WAP. In the example shown theWAP 102 initiates the explicit sounding by sending at time to a nulldata packet announcement (NDPA) frame 240. The NDPA identifies the WAPand the target recipient station(s) for the sounding. Where more thanone station is a target recipient, e.g. an MU-MIMO downlink, the orderin which the recipient stations are listed controls the order of theirfeedback response. Next at time t₁ a null data packet (NDP) 242 is sentby the WAP. This packet like all the other packets associated with thesounding contains no user data rather the header of the packet 242Acontains a ubiquitous preamble field, which in the case of the IEEE802.11ac standard is identified as the VHT-LTF field 266 shown in FIG.2D. The VHT-LTF field a.k.a. channel estimation or sounding field,contains a long training sequence used for MIMO channel estimation bythe receiver. The first intended target e.g. station 108 in the MU-MIMOgroup, then responds at time t₂ with the beamforming feedback packet 244containing communication channel feedback. If the receiving station isIEEE 802.11n compliant the feedback can be in any one of three formats,the link channel matrix H being one of them. If the receiving station isIEEE 802.11ac compliant the feedback is in the form of the actualunitary beamsteering matrix V and the per tone diagonal matrix SNR. Anyremaining stations targeted by the initial sounding, respond with thecommunication channel feedback for their own link when asked to do so bythe WAP. The WAP sends a report poll packet 246 at time t₃ identifyingthe next station from which sounding feedback is requested. Thatstation, e.g. station 112, then responds at time t₄ with beamformingfeedback packet 248. Next, at time is the communication of user datapackets resumes and user data, e.g. MU-MIMO packet 250, is sent on thelink(s) that have been sounded using the associatedprecoding/beamforming matrix determined during the preceding sounding.

FIG. 2D a Prior Art packet diagram of a WLAN packets including thesounding field. All WLAN packets whether associated with communicating asounding or with the communication of user data include a ubiquitousheader portion. All WLAN packet headers include various preamble fieldswith known sequences which allow the receiving station to synchronizereception with packet boundaries and to determine the received channel.What makes a sounding packet a sounding packet is not the sounding fieldin the header, rather the NDPA payload instructions which identify thereceiving stations which are requested to share their channel analysiswith the transmitter so as to improve its subsequent communications.FIG. 2D shows such a packet 242 and the corresponding symbol interval(SI) required to transmit each field thereof. The header 242A includes alegacy portion containing the L-STF, L-LTF and L-SIG fields and a veryhigh throughput portion containing the VHT-SIGA, VHT-STF, VHT-LTF andVHT-SIGB fields. The payload portion 242B of a sounding packet containsno user data. The legacy (L), long (LTF) and short (STF) training andsignal (SIG) 260 fields are compatible with stations supporting only theIEEE 802.11n or earlier standards. The remaining signal and trainingfields are intended only for very high throughput, e.g. IEEE 802.11accompliant devices. The VHT-SIGA field 262 contains information on themodulation and coding scheme (MCS) and number of streams of thesounding. The VHT-STF field 264 is used for automatic gain control(AGC). The VHT-LTF field 266, a.k.a. channel estimation or soundingfield, contains a long training sequence used for MIMO channelestimation by the receiver.

All WLAN packets whether associated with communicating a sounding or thecommunication of user data include a similar header portion with thesame training and signal preamble fields with known sequences whichallow the receiving station to synchronize reception with packetboundaries and to determine the received channel.

FIG. 3 is a timeline of a MU-MIMO downlink in accordance with anembodiment of the invention incorporating hybrid spatial mapping usingboth explicit sounding together with crosstalk tracking. The timelineshows a wireless access point (WAP) 300 and associated stations 302A-Bof a residential WLAN during various phases of operation associated withan MU-MIMO downlink in an embodiment of the invention where both the WAPand the stations include complementary capabilities to enable hybridMU-MIMO spatial mapping. These capabilities reduce the frequency ofexplicit soundings by extending the post sounding interval over which areliable downlink can be maintained. Each packet in the MU-MIMO downlinkhas a payload portion which contains distinct blocks of data eachdestined for a corresponding one of the stations in the MU-MIMO downlinkgroup. The MU-MIMO packet allows transmission of discrete payload datato two or more receiving stations at the same time, thus increasing WLANthroughput. Ideally, each station in the group receives a common MU-MIMOpacket header, and its own a distinct block of data. The distinct blocksof data in the MU-MIMO packet payload are said to be spatially mappedduring transmission so as to arrive at each corresponding one of thestations, without interference, a.k.a. crosstalk, from the distinctblocks of data destined for other stations in the group. The spatialmapping involves complex signal processing using the multiple antennasof the WAP and a precoding matrix to spatially separate the distinctblocks of data in the MU-MIMO packet payload among the targeted MU-MIMOgroup of stations. The determination of the precode matrix, requiresdetailed channel information feedback from each of the targeted stationsin response to an explicit sounding. The feedback from each station asto its communication channel with the WAP is aggregated by the WAP andused to determine the precode matrix for transmitting subsequent MU-MIMOdownlink packets. The minute changes in the communication channels whichwould otherwise blur the spatial separation shortly after an explicitsounding are corrected for by using crosstalk feedback from one or moreof the targeted stations during the MU-MIMO downlink. This avoidsfrequent explicit soundings and extends the downlink interval betweenexplicit soundings, e.g. by 500%.

The timeline in FIG. 3 includes an overhead portion including thesounding and feedback phases “A” and “B” respectively of an Explicitsounding over a representative interval, e.g. 10 milliseconds. Theexplicit sounding is initiated at time to with the soundingtransmissions 308 by the WAP 100. The sounding is conducted on aselected one of the communication channels available to the WLAN. Thesounding phase is initiated with an NDPA (not shown) which announces thegroup of target stations selected for the MU-MIMO sounding andsubsequent downlink. This announcement packet is followed by an NDP 310.Both the NDPA and the NDP packets are sent using an isotropic RF signalstrength 308. Neither the NDPA nor the NDP contains any user data. Thesepackets are explicitly dedicated to soliciting a sounding of the linkchannel. The header 310A of the NDP contains the VHT-LTF training fieldwhich the receiving stations targeted for the MU-MIMO downlink, i.e.stations 302A-B use to characterize the respective link channels H_(a)and H_(b). Each receiving station 302A-B determines the respective linkchannel H_(n), where “n” is the sub-channel or tone index, for each ofthe OFDM tones or sub-channels of the selected channel. The group ofstations 302A-B targeted for the MU-MIMO downlink will then over theinterval t₁-t₂, send the respective channel feedback packets 314, 312including in the payload portion thereof, the channel feedback. In anembodiment of the invention the sounding feedback conforms to the IEEE802.11n standard, where each feedback packet 312, 314 will include the Hmatrix 312B, 314B respectively for each sub-channel or tone of thecorresponding WAP-station sounding. The WAP 300 aggregates this receivedfeedback and determines a precoding matrix “Q” 318A therefrom.

The next portion of the prior art MU-MIMO timeline shown in FIG. 3 isthe extended downlink portion including the downlink transmission andreceipt acknowledgement phases “C, E” and “D, F” respectively of thishybrid spatially mapped MU-MIMO downlink over a representative extendedinterval, e.g. 500 milliseconds. The extension of the downlink intervalfrom 90 to 500 milliseconds is made possible because of the crosstalktracking that takes place during the packet receipt and acknowledgementof the downlink. Initially the precoding matrix “Q” 318A determined fromthe explicit sounding is starting at time t₂, used for spatially mappingthe transmitted MU-MIMO packet 324 to the stations 302A-B. The MU-MIMOpacket 324 includes a header 324A and a payload portion. The spatialmapping of the payload portion of the MU-MIMO packet 324 allows thedistinct block of data for station 302A and the distinct block of datafor station 302B to arrive at their respective target stations withoutcrosstalk or interference from the other block of data. The precodingapplied by the spatial mapper to the transmitted payload produces thedistinctly anisotropic RF signal strength multi-lobe pattern 320A-320B.At time t₃ each station has received the first MU-MIMO packet 324. Eachstation uses the training field in the header thereof, e.g. VHT-LTF toquantify any crosstalk introduced by minute changes in the respectivecommunication channels since the explicit sounding. This crosstalk isquantified by each receiving station in terms of the distinct crosstalkcontribution for each of the other stations in the group. Each station302A-B then acknowledges receipt of the first MU-MIMO packet 324, in an“ACK” packet. In this embodiment of the invention each “ACK” packetincludes in a payload portion thereof, the quantified crosstalk asdetermined by the corresponding station.

At time t₄ the WAP aggregates the crosstalk feedback from the receivedACK packets and uses that crosstalk feedback to determine incrementaladjustments to elements of the precode matrix “Q” in a direction whichreduces the amount of crosstalk at the receiving stations. The WAP thenspatially maps the next transmitted MU-MIMO downlink communicationpacket(s) 344 with the adjusted precode matrix “Q′” 318B and theadjusted anisotropic RF signal strength pattern 340A-B. The receivingstations then repeat the process, acknowledging the receipt of thesecond MU-MIMO downlink packet in ACK packets 350, 352 each of whichincludes the quantified crosstalk as observed by each of the stations,from remaining stations in the group. The downlink and crosstalktracking process repeats itself until, for example, the observed packeterror rate (PER) exceeds an acceptable threshold limit/level, or thesignal-to-interference-plus-noise-ratio (SINR) falls below an acceptablethreshold limit/level at which time, another explicit sounding is made.The advantage of this hybrid approach to spatial mapping incorporatingboth explicit sounding and crosstalk tracking is that only 2 explicitsoundings per second, may be required. Thus overhead is dramaticallyreduced and the downlink portion of MU-MIMO activity is greatlyextended, e.g. 500 milliseconds in the example shown in FIG. 3, vs. theprior art case without crosstalk tracking of 90 milliseconds.

In an embodiment of the invention, some of the benefits of crosstalktracking can be realized even where less than all the stations in agiven MU-MIMO downlink group support crosstalk tracking. In stillanother embodiment of the invention, crosstalk feedback may be carriedin other types of uplink packets, with however some correspondingpenalty in terms of overall overhead.

FIG. 4A is a graph of communication channel variations as a function oftime. MU-MIMO precoding attempts to find a precoding matrix such thattransmissions intended for a given user cause no interference for any ofthe other users included in the MU transmission. For the purposes of thediscussion, we can assume that zero-forcing precoding is used to achievethis interference cancellation.

The precoding is based on knowledge of all the channels between the APand each of the intended receivers. Denote the channel between the APand user i as H_(i). If there are N_(u) users, we have:H _(i) ,i=1, . . . ,N _(u)  (1)where H_(i) is an N_(RX,i)×N_(TX) matrix, assuming that the receiver ihas N_(RX,i) receive antennas and the AP has N_(TX) transmit antennas.For each of the users, the AP determines a precoding matrix Q_(j), j=1,. . . , N_(u), such that:H _(i) Q _(j)=0,i≠j  (2)if the number of streams sent to user j is N_(STS,j), Q_(j) is anN_(TX)×N_(STS,j) matrix.

If we build a composite channel matrix consisting of all individualchannel matrices H_(i) and a full precoding matrix that has the variousQ_(j) as columns, we find that:

$\begin{matrix}{{\begin{bmatrix}H_{1} \\\vdots \\H_{N_{u}}\end{bmatrix}\begin{bmatrix}Q_{1} & \ldots & Q_{N_{u}}\end{bmatrix}} = \begin{bmatrix}{H_{1}Q_{1}} & 0 & 0 \\0 & \ddots & 0 \\0 & 0 & {H_{N_{u}}Q_{N_{u}}}\end{bmatrix}} & (3)\end{matrix}$The entries H_(i)Q_(i) are the effective channels for each of the usersi. The zeros in the matrix indicate that there is no interference fromany of the other users. Residual interference would manifest itself asnon-zero entries for off-diagonal elements. (3) is a (block-)diagonalmatrix of dimensions

$\sum\limits_{i = 1}^{N_{u}}{N_{{RX},i} \times {\sum\limits_{j = 1}^{N_{u}}{N_{{STS},j}.}}}$

The precoding matrices are determined to meet condition (2). However,that calculation is based on a snapshot in time. The channel informationis only really known when the channel is sounded. Any drift in thechannel after that is not captured in the precoding matrices ascalculated. These changes in channel will ultimately lead to a loss ofthe orthogonality condition (2) and the emergence of interference withinthe MU transmission. We will illustrate the effect, and then discussways of countering these changes in channel.

First, we need to find a convenient way to model the changes in channelover time. In the description above, the matrix calculation andproperties apply to a single tone in an OFDM system. For simplicity, wewill continue to express the discussion in matrix framework. In a realOFDM system, every tone can be considered independently. To model thevariation in a given channel, we choose an initial value and a finalvalue for the channel matrix and linearly “morph” from the initial tothe final channel over a selected period of time. This allows us toassess the quality of precoding for a changing channel. The approach isillustrated in FIG. 4A. The first channel realization is a 1×4 matrixshown on the left by circles. The final channel realization is shown onthe right by squares. The lines indicate what the value of the channelis at any intermediate point in time. A similar approach can be used forhigh-dimensional matrices and complex-valued matrices.

FIG. 4A shows two 1×4 channels and their intermediate realizations asthe channel moves from its initial to its final state. Using thisapproach for changing channels, the simulation tries to mimic thetypical behavior of a MU system. Specifically: The system performs achannel sounding at an initial time and derives MU precoding matrices“Q” based on the channel state at this time. These precoding matricesare used for subsequent transmissions, even as the channel changescontinuously. At any time, we can calculate the SINR from the knowledgeof the channel and the precoding matrix that are in use at that time.The signal strength for user i can be calculated as:S _(i)=trace(Q _(i) ⁺ H _(i) ⁺ H _(i) Q _(i))  (4)While the interference experienced by user i can be calculated as:

$\begin{matrix}{I_{i} = {\sum\limits_{j \in i}{{trace}\left( {Q_{j}^{+}H_{i}^{+}H_{i}Q_{j}} \right)}}} & (5)\end{matrix}$The SINR for user i is then given by:

$\begin{matrix}{{SINR}_{i} = \frac{S_{i}}{I_{i} + N_{0}}} & (5)\end{matrix}$where N₀ is power of the additive noise. Throughout the simulations,we'll compare the results with “ideal” precoding. With this, we mean aprecoding where the precoding matrices are always up to date with thelatest channel information. This would be equivalent to performing asounding right before the transmission of the MU packet. Note that therewill still be a small amount of change since there is a small time lagbetween the sounding and the actual transmission, but it is minimal. Thechannels used in the simulation are complex-valued Gaussian channels.

FIGS. 4B-D are graphs of communication channel variations as a functionof time with and without explicit soundings. FIG. 4B shows a case withthree single-antenna users and N_(TX)=4. The precoding matrix iscalculated at time t=0 and not updated after that. We see a quickdrop-off of the SINR as the precoding matrix “ages” and no longermatches the actual channel conditions. FIG. 4B shows the SINR for threeusers as a function of time when the channel changes continuously. Asseen in FIG. 4B, there is a quick drop in SINR as the channel changesaway from the initial value that was used to calculate the precodingmatrices. After a while, the precoding becomes unusable and the systemshould switch from MU to SU to maintain performance. FIG. 4B clearlyillustrates that the precoding matrix can not be used indefinitely in achanging channel. In practice, this means that the AP has to update thechannel information at regular intervals. There are a number ofstrategies to do so. The first and most basic approach is for the AP toregularly sound the channel and re-calculate the precoding matricesbased on the latest channel information.

FIG. 4C shows the SINR of the three users for a case where the AP soundsthe channel ever 100 msec. FIG. 4C shows the SINR for three users as afunction of time when the AP periodically sounds the channel. Thisapproach of periodic sounding improves the situation, but there canstill be drops of about 10 dB relative to the “Ideal” SINR. Note thatright after every sounding, the SINR jumps back to the “ideal”precoding, as expected. The performance goes down steadily after thathowever, leading to a saw-tooth behavior of the SINR over time.

Different propagation environments may need different sounding intervalsfor good performance, so it's hard to find a single sounding intervalfor all environments. A compromise value may be too long for someenvironments and too short for others. In both cases, this results in aloss in performance compared to an ideal sounding frequency. Instead ofsounding at a regular interval, the AP could also use a threshold on theallowed degradation of the original SINR (i.e. relative to the originalchannel and precoding matrix).

FIG. 4D shows a situation where the channel information is refreshedwhen the performance of any one of the users has degraded by more than 3dB. FIG. 4D shows the SINR for three users as a function of time whenthe AP sounds the channel based on observed SINR degradation. FIG. 4Cshows that it is possible with the current sounding protocol to keep thedegradation to within a given range, but at the expense of very frequentchannel sounding. Each channel sounding represents protocol overheadthat reduces the effective throughput.

Another question is how the AP can learn about EVM degradation at thereceiver, since this is not something that can be directly observed.Note however that the receiver has a direct view of the level ofinterference suppression. The current channel estimation protocol ine.g. IEEE 802.11ac allows each receiver to estimate the channels betweeneach stream to its receive antennas, including channels that correspondto streams intended for other users. If (2) is true, the estimatedchannels for these “other” streams should be zero. If the precoding isno longer perfectly orthogonal to the actual channel, the receiver canexplicitly estimate the values of H_(i)Q_(j). If each receiver providesan indication of these values to the transmitter, the AP can effectivelydetermine the right time to sound based on a more accurate picture ofhow the precoding is performing. Note that rate adaptation alone may notbe able to provide such an accurate picture, since even the performanceunder “ideal” precoding is a function of time. Simply observing adegradation in performance is not necessarily an indication that theprecoding is no longer adequate. The feedback from the receiver couldtake a number of different forms: We might consider full channelinformation of H_(i)Q_(j). The information can quantized in some form,since we expect the number to be relatively small. The information couldbe provided on a subset of the tones, since we expect similar behavioron adjacent tones. The information could simply exist of a binaryindication that some threshold has been exceeded. Each of these wouldprovide useful guidance to the AP in determining the right time torefresh channel information. It would avoid situations where thesounding is either too fast or too slow and thereby minimize protocoloverhead, leading to more optimal throughput numbers.

FIG. 4E is a graph of explicit sounding overhead as a function ofmodulation and code schema (MCS) used to transmit sounding feedback.FIG. 4E shows representative results for an 8×8 WAP regularly sounding 64×4 stations on all tones across an 80 MHz channel. This represent thetotal airtime needed per sounding. FIG. 4E shows, it is possible to keepthe SINR degradation within a desired range through a combination ofreceiver feedback and frequent channel sounding. The number of soundingsrequired to keep the degradation to within 3 dB is very high however(about 30 in the example shown in FIG. 4D).

In this section, we explore a different approach to keeping theprecoding matrix “Q” more closely matched to the actual channel—even ifit was originally calculated on a different initial channel. Theapproach does not consist of sounding and re-calculating the precodingmatrix. Instead, we make small incremental changes directly to theprecoding matrix, without even having full information about the channelchange. Let's denote to composite channel consisting of all individualchannels as:

$\begin{matrix}{\overset{\_}{H} = \begin{bmatrix}H_{1} \\\vdots \\H_{N_{u}}\end{bmatrix}} & (6)\end{matrix}$The channel matrix H is the one used to determine the initial precodingcoefficients. However, the channel H will likely change over the timeperiod in which frames are being sent with this particular precodingmatrix. Changes will be gradual over time, so we can express the changeas follows:

$\begin{matrix}{{\begin{bmatrix}H_{1} \\\vdots \\H_{N_{u}}\end{bmatrix}->\begin{bmatrix}{\overset{\sim}{H}}_{1} \\\vdots \\{\overset{\sim}{H}}_{N_{u}}\end{bmatrix}} = {\begin{bmatrix}H_{1} \\\vdots \\H_{N_{u}}\end{bmatrix} + {ɛ\begin{bmatrix}{\Delta\; H_{1}} \\\vdots \\{\Delta\; H_{N_{u}}}\end{bmatrix}}}} & (7)\end{matrix}$i.e. H_(i) changes to {tilde over (H)}_(i), but the change is small whenthe time between the original measurement and the time the packet issent is small enough. This is reflected in the assumption that E is asmall value.

With this change, the matrix (3), which contains all direct and allinterference channels changes from (3) to:

$\begin{matrix}{{\begin{bmatrix}{\overset{\sim}{H}}_{1} \\\vdots \\{\overset{\sim}{H}}_{N_{u}}\end{bmatrix}\begin{bmatrix}Q_{1} & \ldots & Q_{N_{u}}\end{bmatrix}} = \begin{bmatrix}{{\overset{\sim}{H}}_{1}Q_{1}} & {{\overset{\sim}{H}}_{1}Q_{2}} & \ldots & {{\overset{\sim}{H}}_{1}Q_{N_{u}}} \\{{\overset{\sim}{H}}_{2}Q_{1}} & {{\overset{\sim}{H}}_{2}Q_{2}} & \ldots & {{\overset{\sim}{H}}_{2}Q_{N_{u}}} \\\vdots & \vdots & \ddots & \vdots \\{{\overset{\sim}{H}}_{N_{u}}Q_{1}} & {{\overset{\sim}{H}}_{N_{u}}Q_{2}} & \ldots & {{\overset{\sim}{H}}_{N_{u}}Q_{N_{u}}}\end{bmatrix}} & (8)\end{matrix}$In this case, in general: {tilde over (H)}_(i)Q_(j)*0, for i≠j, and{tilde over (H)}_(i)Q_(j)=εΔH_(i)Q_(j). This can be written as:

$\begin{matrix}{{\begin{bmatrix}{\overset{\sim}{H}}_{1} \\\vdots \\{\overset{\sim}{H}}_{N_{u}}\end{bmatrix}\begin{bmatrix}Q_{1} & \ldots & Q_{N_{u}}\end{bmatrix}} = \left. \underset{\underset{diagonal}{︸}}{\begin{bmatrix}{{\overset{\sim}{H}}_{1}Q_{1}} & 0 & 0 \\0 & \ddots & 0 \\0 & 0 & {{\overset{\sim}{H}}_{N_{u}}Q_{N_{u}}}\end{bmatrix}} \middle| {ɛ\underset{\underset{{zero}\mspace{14mu}{on}\mspace{14mu}{diagonal}}{︸}}{\begin{bmatrix}0 & {\Delta\; H_{1}Q_{2}} & \ldots & {\Delta\; H_{1}Q_{N_{u}}} \\{\Delta\; H_{2}Q_{1}} & 0 & \; & {\Delta\; H_{2}Q_{N_{u}}} \\\vdots & \; & \ddots & \; \\{\Delta\; H_{N_{u}}Q_{1}} & {\Delta\; H_{N_{u}}Q_{2}} & \; & 0\end{bmatrix}}} \right.} & (9)\end{matrix}$Non-diagonal elements in the second term represent unwantedinterference. Note that the direct channels in the first term (diagonalmatrix) have also changed with an unknown amount. As we discussedearlier, at receiver “j”, the channel from all streams to the receiveantennas of receiver j can be measured during VHT-LTF (for the case of11ac). This means that this receiver has direct knowledge of allεΔH_(i)Q_(j), i≠j. Let's assume that this information can be conveyed(with some delay, obviously) to the transmitter. When it receives theinformation from all receivers, the transmitter knows the full secondterm in the right-had side of expression (9).

The goal is now to determine an update to the precoding matrix that cancounteract the change in the channel, such that interferencecancellation is maintained or improved. Given that the changes inchannel are assumed to be small (of order ε), we can look for acorrection to the precoding matrix that is of the same order, i.e. Q_(j)changes to {tilde over (Q)}_(j) as follows:[Q ₁ . . . Q _(N) _(u) ]→[{tilde over (Q)} ₁ . . . Q _(N) _(u) ]+ε[ΔQ ₁. . . ΔQ _(N) _(u) ]  (10)If this change of precoding matrix is applied to the changed channel, weget:

$\begin{matrix}{{\begin{bmatrix}{\overset{\sim}{H}}_{1} \\\vdots \\{\overset{\sim}{H}}_{N_{u}}\end{bmatrix}\begin{bmatrix}Q_{1} & \ldots & Q_{N_{u}}\end{bmatrix}} = {\begin{bmatrix}{{\overset{\sim}{H}}_{1}Q_{1}} & 0 & 0 \\0 & \ddots & 0 \\0 & 0 & {{\overset{\sim}{H}}_{N_{u}}Q_{N_{u}}}\end{bmatrix} + {ɛ\begin{bmatrix}0 & {\Delta\; H_{1}Q_{2}} & \ldots & {\Delta\; H_{1}Q_{N_{u}}} \\{\Delta\; H_{2}Q_{1}} & 0 & \; & {\Delta\; H_{2}Q_{N_{u}}} \\\vdots & \; & \ddots & \; \\{\Delta\; H_{N_{u}}Q_{1}} & {\Delta\; H_{N_{u}}Q_{2}} & \; & 0\end{bmatrix}} + {{ɛ\begin{bmatrix}H_{1} \\\vdots \\H_{N_{u}}\end{bmatrix}}\begin{bmatrix}{\Delta\; Q_{1}} & \ldots & {\Delta\; Q_{N_{u}}}\end{bmatrix}} + {\theta\left( ɛ^{2} \right)}}} & (11)\end{matrix}$In the expression (11), we ignore terms of order ε², since they are atleast an order of magnitude smaller than the interference we are tryingto cancel. We now can choose the correction to the precoding matrix,such that terms of order E are eliminated. In other words, we have tosolve the following equations for unknowns ΔQ_(j):

$\begin{matrix}{{{ɛ\begin{bmatrix}0 & {\Delta\; H_{1}Q_{2}} & \ldots & {\Delta\; H_{1}Q_{N_{u}}} \\{\Delta\; H_{2}Q_{1}} & 0 & \; & {\Delta\; H_{2}Q_{N_{u}}} \\\vdots & \; & \ddots & \; \\{\Delta\; H_{N_{u}}Q_{1}} & {\Delta\; H_{N_{u}}Q_{2}} & \; & 0\end{bmatrix}} + {{ɛ\begin{bmatrix}H_{1} \\\vdots \\H_{N_{u}}\end{bmatrix}}\begin{bmatrix}{\Delta\; Q_{1}} & \ldots & {\Delta\; Q_{N_{u}}}\end{bmatrix}}} = 0} & (12)\end{matrix}$Let's denote:

$\begin{matrix}{M_{I} = {ɛ\begin{bmatrix}0 & {\Delta\; H_{1}Q_{2}} & \ldots & {\Delta\; H_{1}Q_{N_{u}}} \\{\Delta\; H_{2}Q_{1}} & 0 & \; & {\Delta\; H_{2}Q_{N_{u}}} \\\vdots & \; & \ddots & \; \\{\Delta\; H_{N_{u}}Q_{1}} & {\Delta\; H_{N_{u}}Q_{2}} & \; & 0\end{bmatrix}}} & (13)\end{matrix}$

With (6) and (13), a solution for ΔQ_(j), i.e. the adjustments to theprecode matrix can be expressed as:ε[ΔQ ₁ . . . ΔQ _(N) _(u) ]=−H ⁺( H H ⁺)⁻¹ M ₁  (14)where −H ⁺(H H ⁺)⁻¹ is identified as the pseudo inverse of the aggregatechannel and M₁ is identified as the interference or crosstalk matrix. Wecan make some important observations about this solution: The solutiononly depends on prior channel information, i.e. the channel H as it wasduring the latest sounding. There is no need to know the updated channel(as in (7)). This matrix can be stored and reused until the nextsounding. The crosstalk matrix M₁ consists entirely of the interferencechannels as measured at the respective receivers. This information isalready implicitly available in the current 11ac training sequence. Theonly new requirement is that this information has to be communicatedback to the MU transmitter.

Once the incremental adjustments to the precode matrix have beendetermined (14), by calculating the product of the pseudo channelinverse of the aggregate channel and the crosstalk matrix, the adjustedprecode matrix “Q′” can simply be calculated by adding the incrementaladjustments to the prior precode matrix as follows:[Q ₁ . . . Q _(N) _(u) ]→[{tilde over (Q)} ₁ . . . {tilde over (Q)} _(N)_(u) ]+ε[ΔQ ₁ . . . ΔQ _(N) _(u) ]Using this adjusted precode matrix “Q′” for spatial mapping ofsubsequently transmitted MU-MIMO downlink packets will reduce thecrosstalk/interference from a magnitude of θ(ε) to a magnitude of θ(ε²).In dB scale, it will double the signal-to-interference ratio (e.g. 20 dBgoes to 40 dB). Below, we show some results that use this updatedprecoding matrix approach.

FIGS. 4F-G are graphs of communication channel variations as a functionof time with the hybrid spatial mapping of an embodiment of theinvention. FIG. 4F shows the SINR of three users when the precodingmatrix is updated based on observed interference channels at therespective receivers. FIG. 4F shows the case where the channel soundingand precoding calculation are performed at time t=0. There is no furthersounding after that, exactly as it was in FIG. 4B. However, theprecoding matrix is updated as explained in the previous section, basedon feedback from the iriterterence channels (13) measured at thereceiver. FIG. 4F should be compared with FIG. 4B. The precodingmatrices keep their cancellation properties for a much longer time. Evenif there is increased degradation towards the end, the performance nevercollapses as it does in FIG. 4B. The degradation in FIG. 4F does howeversometimes exceeds 3 dB, so we could supplement the updating of theprecoding matrix with an occasional sounding. If we set the same 3 dBthreshold, we get the picture as shown in FIG. 4G.

FIG. 4G shows the SINR of three users when the precoding matrix isupdated based on observed interference channels at the respectivereceivers, supplemented with an explicit sounding. FIG. 4G shows thatthe sounding can be substantially reduced when the precoding matrix canbe kept more up-to-date with other means. In FIG. 4G, a single soundingis needed to keep the degradation to within 3 dB for the whole duration.In the equivalent FIG. 4D, about 30 soundings are needed to achieve thesame. And even then, the SINR in FIG. 4D shows an undesirable“saw-tooth” behavior, while the SINR in FIG. 4G is much smoother as afunction of time. The approach of updating the precoding matrices basedon crosstalk/interference channels measured at the receiver shows a lotof promise as an alternative to frequent sounding. It may not replaceexplicit sounding, but when used in combination with explicit soundingit will significantly reduce the needs for explicit channel sounding.

The purpose of hybrid spatial mapping using a mix of explicit soundingand crosstalk tracking is to reduce the protocol overhead that isassociated with sounding. Sending such crosstalk feedback in theubiquitous “ACK” packets is one way of achieving this objective.Additionally, the feedback format for the extra information should bekept sufficiently small. Since we expect the numbers to be typicallysmall, the number of bits required to communicate the information may belimited. In addition, further reduction may be obtained by groupingtones. The analysis presented here applies to a single tone. In an OFDMsystem, the method can be applied to each tone individually. We have tobe mindful of the fact that from one transmission to the next, the tonescould show a phase shift if e.g. symbol alignment is not exactly thesame. The feedback protocol needs a way to encode information in a waythat allows the AP to normalize the received information back to areference phase. The analysis presented here uses full channel “H”information. To keep MU-precoding up to date under changing channelconditions, the precoding matrices need to be kept sufficiently up todate. One approach is to regularly sound the channel to refresh thechannel information and the precoding matrices derived from it. This canbe done autonomously by the WAP, or the WAP can receive supportinginformation from the respective receivers that can help its decision inwhen to refresh the channel information. Information provided by thereceivers can be based on reported crosstalk/interference measurements.

In another embodiment of the invention the crosstalk feedback is used todirectly update the precoding coefficient without explicit soundinginformation. The updates are applied as successive corrections to theprecoding matrix. The updates are a function of crosstalk/interferencechannel measurements as observed at the various receivers.

FIG. 5 is a hardware block diagram of a WAP 300 and two stations 302A Bconfigured to support the hybrid spatial mapping for MU-MIMO downlinksof an embodiment of the current invention. The WAP is shown with aplurality of shared and discrete hardware components coupled together toform four transmit and receive chains for MIMO communications over anarray of antenna 512. The WAP includes an integrated circuit 524hardware component which interfaces with the other transmit and receivepath hardware components to provide the WAP's hybrid spatial mappingcapability in accordance with an embodiment of the invention. The WAPsupports discrete communications with a station or concurrent MU-MIMOcommunications with a group of targeted stations. The WAP in thisembodiment of the invention is identified as a 4×4 WAP supporting asmany as 4 discrete communication streams “a”, “b”, “c”, “d” over itsarray of four antennas 512. The WAP includes: a WLAN stage includingbase band components 500A and radio frequency (RF) components 500Bcoupled to the MIMO antenna array 512. The WAP supports one or more IEEE802.11 wireless local area network (WLAN) protocols.

WAP transmission begins in the baseband stage 500A where outgoingcommunications for associated users/stations singly or in multi-usergroups are subject to baseband transmit processing. The basebandcomponents are configurable to conform with the number of communicationdata streams to be transmitted, which in the example shown is fromone-to-four. The WAP is capable of transmitting up to four separatestreams of data concurrently. The data is packetized in up to fourpacket assemblers 502A-D; encoded for error correction, scrambled,interleaved, and mapped to the appropriate constellation points in up tofour encoder mappers 504A-D, and sent to the spatial mapper 506 forspatial mapping using a precode matrix. In the example shown two streamsof discrete data are subject to baseband processing. Stream “a” destinedfor station 302A and stream “b” destined for station 302B. The spatiallymapped streams “ab” are output by the spatial mapper onto the OFDMtone/sub-channel bins 508A-D at the input of each of the four inversediscrete Fourier transform (IDFT) components 510A-D which transform theoutgoing communications from the frequency to the time domain. There isa discrete precoding matrix for each of the OFDM tones or sub-channels.The outgoing communication data is then passed to the RF stage 500B forupconversion to the appropriate communication channel and transmissionby the four antenna array 512. The spatial mapping using a precodematrix derived from a prior isotropic explicit sounding of the stations,has the effect of spatially separating the distinct “ab” streamstransmitted by the WAP so that upon arrival at stations 302A and 302Bthe “a” stream portion of the “ab” transmission arrives at station 302Awithout crosstalk/interference from the “b” stream, and conversely the“b” stream portion of the “ab” transmission arrives at station 302Bwithout crosstalk/interference from the “a” stream. This spatialseparation is graphically represented by the multi lobed RF signalstrength pattern 512A-B output by the MIMO antenna array 512. Thecomplex signal processing responsible for this result quickly looses itseffectiveness as slight and normally occurring changes in the multi-pathcommunication channel between the WAP and each station take place. Thehybrid spatial mapping capability of the WAP and stations allows postsounding adjustments to the precode matrix which reduce the severity ofthis problem, by using crosstalk feedback during downlink transmissionsto incrementally adjust the precode matrix used for spatial mapping.

WAP reception begins in the RF Stage 500B where uplink communicationsfrom one of the stations are received on each of the four antenna 512.These received communications are downconverted and supplied as input tothe baseband stage 500A. In the baseband stage the receivedcommunications are then transformed from the time to the frequencydomain in the discrete Fourier Transform (DFT) modules 514A-D from whichthey are output as discrete orthogonal frequency division multiplexed(OFDM) tones/sub-carriers/sub-channels 516A-D. The header portion ofeach received uplink packet, and specifically the training fieldtherein, e.g. the VHT-LTF field, is used by the channel estimator 518Ato estimate the frequency dependent distortions in the uplink channeland the updated channel information is passed to the equalizer 518B tocancel out any frequency dependent group or phase delay between thereceived OFDM tones/subchannels. The received communication streams arethen demapped from constellation points to bits, deinterleaved,descrambled and decoded in the corresponding one of up to four demapperdecoders 520A-D and then passed to the corresponding one of up to fourpacket disassemblers 522A-D, which output the received uplinkcommunication.

WAP hybrid spatial mapping is controlled by an integral integratedcircuit 524, and specifically the hybrid spatial mapping circuitry 526thereof which is coupled to the above discussed transmit and receivepath components of the WAP. The explicit channel sounding circuit 528controls the transmission of sounding packets initialized by the packetassemblers 502A-D and the receipt of explicit sounding feedback from thepacket disassemblers 522A-D. The explicit sounding circuit calculatesthe precoding matrix from the explicit sounding feedback and injectsthat matrix into the spatial mapper 506 for spatially mapping single ormulti-user downlink transmissions. For MU-MIMO downlinks, the explicitchannel sounding circuit determines from the received communicationchannel feedback, a MU-MIMO precode matrix “Q” for spatially mapping thetransmission of discrete portions of a payload of the MU-MIMO downlinkcommunication packet for crosstalk-free reception at each of the groupof the associated stations, e.g. stations 302A-B, targeted by theMU-MIMO downlink. The crosstalk damping circuit 530 receives crosstalkfeedback from one or more of the stations in the MU-MIMO downlink groupvia its coupling with the packet disassemblers 522A-D. It determines anyincremental adjustments to the precode matrix on the basis of thecrosstalk feedback. Any adjustments of elements to the precode matrixare in a direction which reduces the amount of crosstalk at thereceiving stations. The adjusted precode matrix is then input to thespatial mapper 506 for spatially mapping subsequently transmittedMU-MIMO downlink communication packets with the adjusted precode matrixthereby improving downlink communications between explicit soundings.

In an embodiment of the invention the hybrid spatial mapping circuit mayinclude a mix of both dedicated and programmable hardware circuitcomponents. These embodiments of the invention may include anon-volatile memory 532 in which any one or all of: program code 534,precode matrices 536A, and pseudo inverse aggregate channel matrices maybe stored. The program code operable on any such programmable portionsof the hardware circuit to perform one or more of the discussedfunctions associated with either or both the explicit channel soundingand crosstalk damping circuits.

Stations eligible for an MU-MIMO downlink may have from one or moreantennas. In the example shown the stations each have only one antenna,though that need not be the case. Station reception in station 302Abegins in the RF Stage 572B where downlink communications from the WAPare received on the antenna 570. These received communications aredownconverted and supplied as input to the baseband stage 572A. In thebaseband stage the received communications are then transformed from thetime to the frequency domain in the DFT module 574 from which they areoutput as discrete orthogonal frequency division multiplexed (OFDM)tones/sub-carriers/sub-channels 574A. The header portion of eachreceived downlink packet, and specifically the training field therein,e.g. the VHT-LTF field, is used by the channel estimator 576A toestimate the frequency dependent distortions in the downlink channel andthe updated channel information is passed to the equalizer 576B toremove phase and amplitude distortions caused by the channel. Thereceived communication stream is then demapped from constellation pointsto bits, deinterleaved, descrambled and decoded in the demapper decoder578 and then passed to the packet disassembler 580, which outputs thereceived uplink communication.

Station 302B is also a single antenna single stream device, though thisneed not be the case. It could for example have an array of antenna, andMIMO multi-stream transmit and receive capability and as suchparticipate as an MU-MIMO downlink target with station 302A. Stationreception in station 302B begins in the RF Stage 542B where downlinkcommunications from the WAP are received on the antenna 540. Thesereceived communications are downconverted and supplied as input to thebaseband stage 542A. In the baseband stage the received communicationsare then transformed to the frequency domain in the DFT module 544 fromwhich they are output as discrete OFDM tones 544A. The header portion ofeach received downlink packet is used by the channel estimator 546A toestimate the frequency dependent distortions in the downlink channel andthe updated channel information is passed to the equalizer 546B toequalize the received OFDM tones. The received communication stream isthen demapped, deinterleaved, descrambled and decoded in the demapperdecoder 548 and then passed to the packet disassembler 550, whichoutputs the received uplink communication.

Station transmission of station 302A begins in the baseband stage 572Awhere outgoing uplink communications to the WAP are subject to basebandtransmit processing. The station transmits a single stream of data. Thedata is packetized in the packet assembler 582; encoded for errorcorrection, scrambled, interleaved, and mapped to the appropriateconstellation points in the encoder mapper 584, and sent to the spatialmapper 586 for any spatial mapping. In the instant example, where thestation has only one antenna no spatial mapping is possible. Theoutgoing communication stream is then passed onto the OFDMtone/sub-channel bins 588A at the input of the IDFT component 588 whichtransform the outgoing communications from the frequency to the timedomain. The outgoing communication data is then passed to the RF stage572B for upconversion to the appropriate communication channel andtransmission by the antenna 570.

Station transmission of station 302B begins in the baseband stage 542Awhere outgoing uplink communications to the WAP are subject to basebandtransmit processing. The station transmits a single stream of data. Thedata is packetized in the packet assembler 552; encoded, scrambled,interleaved, and mapped to the appropriate constellation points in theencoder mapper 554, and sent to the spatial mapper 556 for any spatialmapping. In the instant example, where the station has only one antennano spatial mapping is possible. The outgoing communication stream isthen passed onto the OFDM tone/sub-channel bins 558A at the input of theIDFT component 558 which transform the outgoing communications from thefrequency to the time domain. The outgoing communication data is thenpassed to the RF stage 542B for upconversion to the appropriatecommunication channel and transmission by the antenna 540.

Hybrid spatial mapping feedback operations for station 302B arecontrolled by an integral integrated circuit 560. Hybrid spatial mappingfeedback for station 302A is controlled by an integral integratedcircuit 590. The integral integrated circuit 560 for station 302B willbe discussed in detail, and is functionally similar to the circuitry(not shown) for the other station. The hybrid spatial mapping feedbackcircuitry 562 is coupled to the above discussed transmit and receivepath components of the station 302B. The channel estimation circuit 564Ais coupled to the channel estimator 546A and is responsive to anexplicit sounding received from the WAP to determine a communicationchannel associated therewith and to transmit feedback to the WAP as tothe determined communication channel via the packet assembly component552 in a sounding feedback packet. The crosstalk tracker circuit 564A isalso coupled to the channel estimator 546A and is responsive to thetraining field, e.g. the VHT-LTF field, in a received downlink MU-MIMOpacket received from the WAP to determine the quantity of crosstalk insaid packet from portions of the downlink MU-MIMO packet payloaddestined for other stations in the MU-MIMO group and to transmitfeedback to the WAP as to the determined crosstalk via the packetassembly component 552 in for example the payload of an “ACK” packet,which both acknowledges receipt of the MU-MIMO downlink and quantifiesthe amount of crosstalk if any, in the received MU-MIMO packet.

In an embodiment of the invention the hybrid spatial mapping feedbackcircuit may include a mix of both dedicated and programmable hardwarecircuit components. These embodiments of the invention may include anon-volatile memory 568 in which program code 569 may be stored. Theprogram code is operable on any such programmable portions of thehardware circuit to perform one or more of the discussed functionsassociated with either or both the channel estimation or crosstalktracking circuits.

FIG. 6 is a detailed circuit diagram of hybrid spatial mapping circuitsin the WAP and Stations as shown in FIG. 5, in accordance with anembodiment of the current invention. On the WAP an embodiment of thehybrid explicit channel sounding and crosstalk damping circuits 600, 630respectively are shown. On the targeted downlink stations an embodimentof the hybrid MU-MIMO feedback circuits, specifically the channelestimation and crosstalk tracking circuits 670/678 and 680/688respectively are shown.

On the WAP the explicit channel sounding circuit 600 initializes anexplicit channel sounding by assembling a NDP sounding packet includingthe training field, e.g. VHT-LTF 624 in the packet assembler 622 andspatially maps the training field using a sounding matrix 602 as inputto the spatial mapper 626. The sounding packet is transmitted, withgenerally isotropic RF signal strength 660, by the MIMO antenna array(not shown) to the two or more receiving stations in the MU-MIMO group.

On each receiving station the integral channel estimator circuit, e.g.channel estimator 670 determines the communication channel 674associated with the received VHT-LTF or other training field 672 of theexplicit sounding and transmits a sounding feedback packet 676 to theWAP's explicit channel sounding circuit 600 of the channel informationderived from the sounding.

On the WAP, the packet disassembler 604 of the explicit channel soundingcircuit 600 unpacks the channel information 606 from all the feedbackpackets from the stations in the MU-MIMO group. Then the WAP performstwo calculations. The first calculation is the determination of aprecode matrix “Q” for the spatial mapping the MU-MIMO downlinkcommunication packet. That calculation may involve a singular valuedecomposition (SVD) 608 of the aggregated channel feedback 606 todetermine the precode matrix “Q”. The second calculation is thedetermination of the pseudo inverse of the aggregate channels (PIAC)610. The explicit channel sounding circuit initiates the transmission ofthe downlink transmission by assembling in the packet assembler 622 theMU-MIMO downlink packet including the VHT-LTF training field as well asthe discrete “a” and “b” streams of data 620 in the payload of theMU-MIMO packet. The MU-MIMO packet including the training and payloadfields is then spatially mapped with the precode matrix “Q” in thespatial mapper 626 and transmitted by the WAP's MIMO antenna array (notshown) to the downlink stations. The downlink MU-MIMO packettransmission exhibits an anisotropic RF signal strength profile 662A-Bso that the distinct “a” and “b” portions of each downlink packet areeach received at the corresponding one of the stations withoutinterference/crosstalk from the portions destined for other stations inthe MU-MIMO group. The explicit channel sounding circuit also outputsthe precode matrix “Q” and PIAC to the crosstalk damping circuit 630.

On each receiving station the integral crosstalk tracker circuit e.g.crosstalk tracker 680 determines the amount of any crosstalk in thereceived packet and specifically in the VHT-LTF or other training field682. Crosstalk is quantized on a per station, e.g. per stream, basis asto the portions of the packet destined for remaining stations in theMU-MIMO group. The quantized crosstalk is then transmitted by thecrosstalk tracker 680 back to the WAP in a feedback packet 686, which inan embodiment of the invention is the “ACK” packet used to acknowledgereceipt of the packet.

On the WAP the crosstalk damping circuit 630 and specifically the packetdisassembler 626 thereof unpacks the received packet and passes thecrosstalk feedback therein, to the aggregator 626. The aggregatorpopulates the appropriate ones of the off diagonal elements of theinterference matrix “M” 636 with the corresponding elements of thecrosstalk feedback aggregated from one or more of the targeted station sin the MU-MIMO group. The crosstalk damping circuit then obtains theproduct 634 of the PIAC 632 obtained from by the explicit channelsounding circuit and the interference matrix 636 and passes theresultant 638 offset matrix 640 to the next circuit stage. The offsetmatrix containing the incremental adjustments to the precode matrix isthen added 642 to either the initially determined precode matrix “Q” orto some subsequently derived variation thereof as determined by themultiplexer 648. The multiplexer has two inputs 648A-B with the “A”input associated with the initial precode matrix “Q” and the “B” inputassociated with subsequent adjustments to same, e.g. “Q′”, “Q″”, “Q′″”,“Q″″” etc. The multiplexer is reset to the “A” input after each explicitsounding and is otherwise on the “B” input. The output of themultiplexer is coupled to the prior precode matrix register 644 thecontents of which are added 642 to the contents of the offset matrixregister 640. The sum 646 represents the adjusted precode matrix whichis input to the spatial mapper for spatially mapping subsequent downlinkMU-MIMO packet transmissions, until such time as on the basis of forexample PER or SINR another explicit sounding takes place.

FIG. 7 is a process flow diagram of processes associated with hybridspatial mapping for MU-MIMO downlinks on both the WAP and stations inaccordance with an embodiment of the invention. Processing begins withthe block of processes 700 associated with the Explicit MU-MIMO channelsounding phase. In the initial process 702 the group of stationsselected for an MU-MIMO downlink receive an explicit sounding from theWAP which comprises a known preamble modulated on the maximum Lumber ofstreams from which each station can provide feedback. The soundingcomprises a known training sequence modulated on the training field,e.g. the VHT-LTF held, of the sounding packet, e.g. the NDP packet. Thenin process 704 the stations each use the training field to determine thechannel state information (CSI), i.e. the respective link channels withthe WAP. Next in process 706 each station wirelessly transmits thesounding feedback to the WAP in the form of the respective “H” matrices.In process 708 the WAP receives the CSI sounding feedback from each ofthe stations targeted for the MU-MIMO downlink. The WAP determines theprecode matrix “Q” from the sounding feedback. The determination of theprecode matrix may be made by performing a singular value decomposition(SVD) of the combined channel matrices “H”. The precode matrix “Q” isthen used to spatially map subsequent. MU-MIMO downlink wirelesstransmissions. The WAP also in process 710 determines the pseudo inverseof the channel matrix, i.e. −H ⁺(H H ⁺)⁻¹ from the feedback and storesthe transform for use during the next phase of hybrid spatial mappingwhich occurs during the downlink wireless transmissions.

The second block of processes 720 is associated with crosstalk dampingduring the MU-MIMO downlink communication phase. In the initial process722 the stations receive the MU-MIMO downlink packets and track thelevel of Crosstalk/interference from the portion of the MU-MIMO packet,i.e. the stream or streams destined for each of the other stations inthe MU-MIMO group. Each station uses the training symbols in thepreamble header field, e.g. the VHT-LTF field in the MU-MIMO downlinkpacket which contains known or pre-defined: modulation, number ofstreams, and bit sequence to analyze changes thereto brought about bythe link channel, e.g. fading, attenuation, and phase shift, and usesthis information to demodulate the downlink MU-MIMO packet.Additionally, each receiving station uses this training field toquantify the crosstalk from the portions of the MU-MIMO packet, i.e. thestreams destined for other stations in the group. Then in process 724the crosstalk information is transmitted back, as crosstalk feedback tothe WAR In an embodiment of the invention, the feedback is included inthe payload portion of the ubiquitous “ACK” packet which each stationuses after receipt of one or more downlink packets to acknowledge to theWAP the receipt of those packets. The WAP in process 726 receive thefeedback from one or more of the downlink MU-MIMO stations and populatescorresponding off-diagonal elements of the interference Matrix “M” withthe crosstalk feedback information. Then in process 728 the WAPcalculates the product of the pseudo inverse of the aggregate channeland the Interference matrix and adds the resultant offset matrix to thecurrent precode matrix to obtain the adjusted precode matrix “Q′”. TheWAP then in process 730 uses the adjusted precode matrix to spatiallymap subsequently transmitted MU-MIMO downlink packets. Next the WAPdetermines some measures of downlink efficiency or throughput, e.g.packet error rate (PER) or signal-to-interference-plus-noise-ratio(SINR). Then in decision process 732 a determination is made as towhether or not to continue downlink transmissions using intermittentadjustments of the precode matrix or to temporarily halt downlink packettransmissions to allow for another explicit sounding sequence. If packeterror rate (PER) rises above a threshold level, orsignal-to-interference-plus-noise-ratio (SINR) for the downlink MU-MIMOcommunication packets falls below a threshold level control may returnto process block 700 for the next explicit sounding. Alternately, if PERand or SINR are still at acceptable levels control may return to thecrosstalk damping block of processes 720 which are repeated therebyextending the interval between explicit soundings. In this latter case,the resultant offset matrix in process 728 is added to the currentprecode matrix, e.g. “Q′” to obtain the twice adjusted precode matrix,e.g. “Q″”.

The components and processes disclosed herein may be implemented asoftware, hardware, firmware, or a combination thereof, withoutdeparting from the scope of the Claimed Invention.

The foregoing description of a preferred embodiment of the invention hasbeen presented for purposes of illustration and description. It is notintended to be exhaustive or to limit the invention to the precise formsdisclosed. Obviously many modifications and variations will be apparentto practitioners skilled in this art. It is intended that the scope ofthe invention be defined by the following claims and their equivalents.

What is claimed is:
 1. A wireless access point (WAP) having a plurality of antennas and supporting a wireless local area network (WLAN) including multiple-input multiple-output (MIMO) communications with associated stations on a selected communication channel, the WAP comprising: hardware processing circuitry to perform hybrid spatial mapping operations for multi-user (MU) MIMO downlinks including: an explicit channel sounding circuit for sounding targeted stations among the associated stations for a multi-user (MI) MIMO downlink and for determining from explicit sounding feedback received from said targeted stations both precode matrix “Q” for spatially mapping the MU-MIMO downlink and a “P1AC” matrix corresponding to a pseudo inverse of an aggregate channel between the WAP and the targeted stations; and a crosstalk damping circuit responsive to crosstalk feedback received from the targeted stations to combine the P1AC matrix with the received crosstalk feedback into an offset matrix and to spatially map subsequent MU-MIMO downlinks with an adjusted precode matrix “Q” de rived by combining the offset matrix with the precode matrix “Q” to improve the MU-MIMO downlink between explicit soundings.
 2. The WAP of claim 1, wherein the crosstalk damping circuit receives the crosstalk feedback in acknowledgment (ACK) packets from the targeted stations to avoid interruption of the MU-MIMO downlink.
 3. The WAP of claim 1, wherein the PIAC matrix determined by the explicit channel sounding circuit corresponds to: −H ⁺(H H ⁺)⁻¹ where H is a channel matrix.
 4. A method for operating a wireless access point (WAP) having a plurality of antennas and supporting a wireless local area network (WLAN) including multiple-input multiple-output (MIMO) communications with associated stations on a selected communication channel, the method comprising acts of: explicitly sounding targeted stations among the associated stations for a multi-user (MU) MIMO downlink; determining from explicit sounding feedback received from said targeted stations both a precode matrix “Q” for spatially mapping the MU MIMO downlink and a “P1AC” matrix corresponding to a pseudo inverse of an aggregate channel between the WAP and targeted stations; transmitting MU-MIMO downlink packets to the targeted stations spatially mapped with precode matrix “Q”; receiving crosstalk feedback from at least one of the targeted stations response to the MU-MIMO downlink packets; combining the P1AC matrix and the received crosstalk feedback into an offset matrix; deriving an adjusted precode matrix “Q′” by additively combining the precode matrix “Q” with the offset matrix; spatially mapping subsequently transmitted MU-MIMO downlink packets to the targeted stations with the adjusted precode matrix “Q′”; and; repeating the receiving, combining, driving, and spatially mapping acts to increase an interval between explicit soundings.
 5. The method for operating a WAP of claim 4, wherein the act of receiving crosstalk feedback further comprises: receiving crosstalk feedback in acknowledgment (ACK) packets from the targeted stations to avoid interruption of the MU-MIMO downlink.
 6. The method for operating a WAP of claim 4, wherein the MAC matrix determined in the determining act corresponds to: −H ⁺(H H ⁺)⁻¹ where H is a channel matrix. 